Noise shapers with shared and independent filters and multiple quantizers and data converters and methods using the same

ABSTRACT

A noise shaper including first and second quantizers and first and second feedback paths each providing feedback from a corresponding quantizer output. A loop filter system implements a plurality of transfer functions including a first non-zero transfer function between the first feedback path and an input of the first quantizer, a second non-zero transfer function between the first feedback path and an input of the second quantizer, a third non-zero transfer function between the second feedback path and the input of the first quantizer and a fourth non-zero transfer between the second feedback path and the input the second quantizer.

BACKGROUND OF THE INVENTION

1. Field of Invention

The present invention relates in general to delta-sigma modulators andin particular, to noise shapers with shared and independent filters andmultiple quantizers and data converters and methods using the same.

2. Background of Invention

Delta-sigma modulators are particularly useful in digital to analog andanalog to digital converters (DACs and ADCs). Using oversampling, thedelta-sigma modulator spreads the quantization noise power across theoversampling frequency band, which is typically much greater than theinput signal bandwidth. Additionally, the delta sigma modulator performsnoise shaping by acting as a highpass filter to the noise; most of thequantization noise power is thereby shifted out of the signal band.

The typical delta sigma modulator in an ADC includes an input summerwhich sums the analog input signal with negative feedback, an analoglinear (loop) filter, a quantizer and a feedback loop with a digital toanalog converter (feedback DAC) coupling the quantizer output and theinverting input of the input summer. A delta-sigma DAC is similar, witha digital input summer, a digital linear filter, a digital feedbackloop, a quantizer and an output DAC at the modulator output. In a firstorder modulator, the linear filter comprises a single integrator stage;the filter in higher order modulators normally includes a cascade of acorresponding number of integrator stages. Higher-order modulators haveimproved quantization noise transfer characteristics over modulators oflower order, but stability becomes a more critical design factor as theorder increases. For a given topology, the quantizer may be either aone-bit or a multiple-bit quantizer.

The feedback DACs in multi-bit delta-sigma ADCs, as well as the outputDACs in multi-bit delta-sigma DACS, are typically constructed fromweighted conversion elements. Each conversion element converts onedigital bit into a weighted-step analog voltage or current. The currentsor voltages generated by the weighted conversion elements for thedigital word being converted are then summed to generate the analogoutput signal. Mismatch between conversion elements, however, causes theweighted steps of current or voltage to deviate from their idealweighted-step values. Element mismatch results in mismatch noise anddistortion in the output signal. Consequently, dynamic element matching(DEM) circuitry is normally included at the DAC inputs which spreads themismatch noise across the analog output signal band.

For example, a number of well-known DEM designs includingbarrel-shifting, individual level averaging, butterfly routing, and dataweighted averaging, exist. DEM circuits however do have significantdrawbacks. Also, in multiple-bit modulators the DEM circuitry isrelatively large, especially in high voltage ADCs requiring a largefabrication geometry. In addition, a tendency for the DEM circuit tobecome tonal exists, and the DEM circuit is typically a low order,delta-sigma modulator.

Hence what is needed are techniques which address the problem ofmismatch between data conversion elements in DACs and ADCs. Suchtechniques should, for example, eliminate or minimize the DEM circuitryrequired in the given DAC or ADC.

SUMMARY OF INVENTION

The principles of the present invention generally apply to noise shaperswith multiple quantizers and shared and independent filter functions. Inone representative embodiment, a noise shaper is disclosed includingfirst and second quantizers and first and second feedback paths eachproviding feedback from a corresponding quantizer output. A loop filtersystem implements a plurality of transfer functions including a firstnon-zero transfer function between the first feedback path and an inputthe first quantizer, a second non-zero transfer function between thefirst feedback path and an input of the second quantizer, a thirdnon-zero transfer function between the second feedback path and theinput of the first quantizer and a fourth non-zero transfer between thesecond feedback path and the input the second quantizer.

Noise shapers embodying the inventive principles have substantialadvantages over the prior art. For example, modulators with both sharedand independent filter stages and multiple-quantizers allow for thecharacterization of both global noise shaping across all modulatoroutputs and local noise shaping at individual modulator outputs. Globalnoise shaping is the ability of the delta-sigma modulator to shape thetotal quantization noise spectrum generated by the sum of the outputspectrums of the multiple quantizers. Local noise shaping is the abilityof the delta-sigma modulator to shape the spectrum of the difference ofthe spectrums output from the multiple quantizers. In other words,global noise shaping characterizes the overall shaping modulator outputspectrum and local noise shaping allows the difference noise spectrumexposed to mismatch in the following conversion elements to be shaped.Generally, an improvement in global noise shaping results in a reductionin the local noise shaping capability, and vice versa. Furthermore, theinventive principles may be applied to a number of different modulatortopologies, including feedforward, feedback, and combinationfeedforward—feedback topologies.

BRIEF DESCRIPTION OF DRAWINGS

For a more complete understanding of the present invention, and theadvantages thereof, reference is now made to the following descriptionstaken in conjunction with the accompanying drawings, in which:

FIG. 1A is a high-level functional block diagram of an exemplarydelta-sigma analog to digital converter (ADC) system embodying theprinciples of the present invention;

FIG. 1B is a high level functional block diagram of an exemplary digitalto analog converter (DAC) system embodying the principles of the presentinvention;

FIG. 2 depicts an exemplary switched capacitor output DAC, which may beutilized in the DAC of FIG. 1B,

FIGS. 3A AND 3B depict an exemplary switched-capacitor feedback DAC,which may be utilized in the DAC of FIG. 1A;

FIGS. 4A and 4B are conceptual diagrams of a delta-sigma DAC withmultiple quantizers, a shared filter for overall noise shaping andindependent filters for mismatch noise shaping according to theinventive principles;

FIG. 5 is a block diagram of an exemplary feed-forward delta-sigmamodulator topology utilizing shared and independent sets of filterstages and multiple quantizers according to the present principles;

FIG. 6 illustrates an exemplary feedback modulator topology with bothshared and independent filters and multiple quantizers according to theinventive principles;

FIG. 7 depicts an exemplary delta sigma modulator topology with sharedand independent filter stages and multiple quantizers, in which theshared filter is fed by the independent filters;

FIG. 8 illustrates a further exemplary feedback delta sigma modulatortopology with shared and independent filters and multiple quantizersembodying the principles of the present invention;

FIG. 9 illustrates an exemplary delta sigma modulator topology withmultiple quantizers which includes a shared filter and multipleindependent filters for noise shaping the difference between four (4)output signals; and

FIG. 10 is a schematic representation of an exemplary dither source(generator) suitable for introducing dither adding up to a constant tothe quantizer inputs of the delta sigma modulators of FIGS. 4-9.

DETAILED DESCRIPTION OF THE INVENTION

The principles of the present invention and their advantages are bestunderstood by referring to the illustrated embodiment depicted in FIGS.1-10 of the drawings, in which like numbers designate like parts.

FIG. 1A is a high-level functional block diagram of an exemplarydelta-sigma analog to digital converter (ADC) system 100 embodying theprinciples of the present invention. ADC 100 is useful in a number ofsignal processing data acquisition, and similar applications requiringthe conversion of analog signals from a given analog source 101 into thedigital domain.

The input analog signals are passed through an analog low-passanti-aliasing filter 102 which removes out-of-band signals and noisethat would otherwise fold back into the signal baseband duringsubsequent modulation. Modulation is performed in a delta-sigmamodulator (noise shaper) 103, which according to the inventiveprinciples includes a set of global (shared) analog filter stages 104, aset of local analog filter stages 105, and multiple quantizers 106. (InFIG. 1A, global filter stages 104 are shown in front of local filterstages 105 and quantizers 106; as will be discussed below, this orderingvaries depending on the specific noise shaper topology). The modulatorfeedback loop includes optional dynamic element matching (DEM) circuitry107, a feedback DAC 108 and an input summer 109. The output frommultiple quantizers 106 are summed and digitally filtered in outputstage 110. Exemplary delta-sigma modulator topologies embodying theinventive principles and suitable for use in analog noise shapers suchas modulator 103 of ADC 100 are discussed in detail below.

The inventive principles are also be embodied in digital delta-sigmamodulation applications such as exemplary digital-to-analog converter(DAC) system 111 shown in FIG. 1B. In DAC 111, digital data from adigital data source 112, such as a compact disk (CD) or digitalversatile disk (DVD) player, are passed through a digital interpolationfilter 113 which increases the sampling rate by a given oversamplingfactor. The upsampled data are then noise shaped by a digitaldelta-sigma modulator 114.

The exemplary modulators discussed below, using digital stages, are alsoapplicable to digital applications such as digital delta-sigma modulator114. Generally, digital delta-sigma modulator 114 includes a set ofglobal (shared) digital filter stages 115, a set of local (independent)digital filter stages 116, and multiple quantizers 117. The digital dataoutput from quantizers 117 are summed by feedback summer 118 whichprovides the negative feedback to modulator input summer 119. (Again, asdiscussed further below, the order of global stages 115, local stages116, and quantizers 117 varies depending on the modulator selected). Theconverter system output stages include optional DEM circuit 120 a and120 b, an output DAC 121, (e.g., a switched-capacitor or currentsteering DAC), and an analog low pass filter 122.

Some non-zero mismatch occurs between the elements of feedback DAC 108,in the case of delta-sigma ADC system 100, or between the elements ofoutput DAC 121, in the case of delta-sigma DAC system 111. For purposesof illustrating element mismatch in feedback DAC 108 and output DAC 121,exemplary electrical schematic diagrams of multiple-bitswitched-capacitor output and feedback DAC 200 and 300 are respectivelyshown in FIGS. 2 and 3. (Element mismatch occurs in other types of DACs,such as current steering DACs, as well; switched-capacitor DACs havebeen chosen for illustrative purposes). An exemplary switched capacitoroutput DAC 200, which may be utilized for DAC 121 of FIG. 1B, is shownin FIG. 2, and an exemplary switched-capacitor feedback DAC 300, whichmay be utilized for DAC 108 of FIG. 1A, is shown in FIGS. 3A and 3B.

FIG. 2 depicts an output DAC 200 operating on quantized digital samplesof n+1 number of bits, in which n is an integer greater than one. Dataconversion on the input bits D0-Dn and the complementary bits /D0-/Dn isperformed by a corresponding set of conversion elements 201. If inputbits D0-Dn and complementary bits /D0-/Dn are thermometer encoded, eachelement 201 has unit weight and includes an input sampling switch 202for sampling charge onto a corresponding sampling capacitor 204 (Cs)during the sampling phase (φ₁) and a second switch 203 for forcingcharge from sampling capacitor 204 during the integration phase (φ₂).Switches 205 couple the opposing plates of sampling capacitors (Cs) 204to the common mode voltage (Vcm) during sampling and switches 206 passcharges from sampling capacitors 204 to the output stage duringintegration. The output stage includes a conventional operationalamplifier 208 and integration capacitors 209 a and 209 b.

Output DAC 200 operates generally as follows. Switches 205 close at thestart of the sampling phase (Phase 1-φ₁), and after a delay (Phase 1delayed-φ_(1D)), input switches 202 close to sample the input bits D0-Dnand the complements /D0-/Dn onto the input plates of sampling capacitors204. Switches 203 and 206 are open during Phase 1. During theintegration phase (Phase 2-φ₂), switches 206 initially close, and aftera delay (Phase 2 delayed-φ_(2D)), switches 203 close to force the chargeon the input plates of sampling capacitors 204 to the correspondingsumming node of opamp 208 and integration capacitor 209 a or 209 b.During Phase 2, switches 202 and 205 are open.

In the ideal case, unit conversion elements 201 would provide charge inequal unit steps to the integrator summing nodes during the integrationphase. However, in actual devices, mismatch between the values of themultiple capacitors 204 will result in variations in the charge stepsgenerated by conversion element 201. A one-percent (1%) mismatch betweenelements 204 approximately increases the overall noise floor to −40 dBrelative to the noise floor of a single output. This noise thendominants the system. Hence, DEM circuitry is typically utilized whichroutes bits D0-Dn and complementary bits /D0-/Dn to differentcombinations of conversion elements 201 such that the utilization ofeach conversion elements is approximately equalized. While this routingspreads the mismatch noise across the output signal band, DEM techniquesalso produce tonality in the output noise, depending on the DEMalgorithm utilized. The DEM circuitry also adds size and complexity tothe design.

Mismatch error must also be addressed in feedback DACs, such as feedbackDAC 108 of ADC 100 shown in FIG. 1A. FIGS. 3A and 3B illustraterepresentative switched-capacitor analog integrator stage/summer 300with multiple-bit DAC feedback from quantizers 106 suitable for use asfeedback DAC 108 of FIG. 1A. As with output DAC 200, an exemplarytwo-phase switch-capacitor design will be described.

For the differential input data paths sampling input signals V_(in+)andV_(in−1) switches 304 a and 304 b close during sampling Phase 1 (φ₁) tocouple the top plates of input sampling capacitors (C_(IN)) 303 a and303 b to the common mode voltage (V_(cm)). During Delayed Phase 1(φ_(1D)), switches 301 a and 301 b close and the differential inputvoltage V_(IN) is sampled onto input plates of sampling capacitors(C_(IN)) 303 a and 303 b. Switches 302 a-302 b and 305 a-305 b are openduring sampling (φ₁).

Also, during the sampling phase (φ₁), the reference voltage V_(REF)(V_(REF+)−V_(REF−)) is sampled through feedback DAC 300. Tworepresentative unit DAC elements 320 of an exemplary DAC operating onn+1 number of bits in response to digital bits D0-Dn and thecomplementary bits /D0-/Dn from quantizers 106 of FIG. 1A are shown infurther detail FIG. 3B. In particular, V_(REF) is sampled onto referencesampling capacitors (C_(REF)) 306 a and 306 b for each bit path 320 byswitches 307 a and 307 b and 304 a and 304 b (as shown in FIG. 3A).Switches 309 a and 309 b (FIG. 3A) are open during the sampling phase.

Switches 310 a-310 d for each path (as shown in FIG. 3B), under thecontrol of complementary bits D0-Dn and /D0-/Dn from the associatedquantizer 106, couple or cross-couple the input plates of referencesampling capacitors C_(REF) 306 a and 306 b to the reference voltageV_(REF) being sampled by reference sampling switches 307 a and 307 b. Inother words, the configuration of switches 310 a-310 d for a givenreference sampling path 320 sets the polarity of the voltage at theinput plates of the corresponding reference sampling capacitors 306 aand 306 b as a function of quantizer feedback.

During the integration phase (φ₂) the switches reverse theirconfiguration with switches 302 a and 302 b closing, and switches 301a-301 b and 304 a-304 b opening for the input signal paths. For thereference paths, switches 307 a and 307 b open and switches 309 a and309 b close. The charges on the input plates of input samplingcapacitors C_(IN), and reference sampling C_(REF) are forced to theoutput (top) plates and charge sharing nodes A and B. During DelayedPhase 2 (φ_(2D)) switches 305 a and 305 b close to transfer the chargeat nodes A and B from the top plates of input and reference samplingcapacitors C_(IN) and C_(REF) to the summing nodes at the inverting (−)and non-inverting (+) inputs of opamp 312 and integrator capacitors (C₁)311 a and 311 b.

With respect to feedback DAC—summer—integrator stage 300, mismatchbetween elements (bit paths) 320 results in non-linearities in thecharge contributions at nodes A and B. DEM circuitry is thereforetypically included in the feedback (DAC) path to equalize elementutilization. The case of a feedback DAC, the DEM circuitry is normallyand significantly large, especially in the case of a high-voltage ADC.

Generally, modulators 103 and 114 respectively of FIGS. 1A and 1B aredesigned by trading off between global (overall) noise shaping and localnoise shaping to account for DAC element mismatch. Specifically, globalfilter stages 104 and 115 in conjunction with the local filter stages105 or 116 set the overall noise shaping characteristics of the givenmodulator 103/114 while local filter stages 105 or 116 set the localnoise shaping characteristics for corresponding sets of DAC elements.The resulting system shapes the overall noise (of the sum of theelements) to a greater extent than the noise at each quantizer output.This operational characteristic is similar to the function provided intwo stages by a DEM. Consequently, the conventional DEM circuitry iseliminated, or at least made significantly simpler and smaller. As anexample, the quantizers may be all single bit, removing the need for anyDEM circuitry. Alternatively, the quantizers could be three-level withsimple two-element DEMs per quantizer.

FIG. 4A is a conceptual (general) representation of a delta-sigma DACtopology 400 with shared and independent filters and multiplequantizers. DAC topology 400 includes a loop filter system 401, twoquantizers 402 a-402 b, two DACs 403 a-403 b and an output summer 404.Quantizers 402 a and 402 b are shown as multiple-bit quantizers and aremodeled as additive quantization noise sources, as generally shown inFIG. 4B. Two feedback loops 405 a and 405 b, which include delays fortiming, couple the outputs X₁ and X₂ of DAC1 403 a and DAC2 403 b backto the inputs of loop filter system 401. While a two quantizer—twofeedback loop DAC topology 400 is shown in FIG. 4A for illustrativepurposes, the concepts described with respects to modulator 400 may beextended to any modulator topology with n number of multiple quantizersand n number of corresponding feedback loops, n being an integer greaterthan one. The general concepts and principles taught by the shared andindependent filters 402 a and 402 b of loop filter system 401 andmultiple quantizers result in the ability to provide global and localnoise shaping for various modulator topologies, such as illustrated inexemplary modulator topologies 500, 600, 700, 800 and 900 discussedbelow.

For the representative two quantizer—two feedback loop embodiment, loopfilter system implements four transfer functions (respectively h₁₁, h₁₂,h₂₁, h₂₂) between the outputs of the two feedback loops 405 a-405 b andthe inputs to the two quantizers 402 a-402 b. Transfer functions h₁₁,h₁₂, h₂₁, h₂₂ are all non-zero functions, and at least two transferfunctions h₁₁, h₁₂, h₂₁, h₂₂ are different. For purposes of describingapplication of the inventive principles, h₁₁=h₂₂ and h₁₂=h₂₁. However,these relationships are not a strict requirement for practicing theinventive principles.

With h₁₁=h₂₂ and h₁₂=h₂₁, the noise at the output of summer 404 (Output)is: $\begin{matrix}{{\frac{n_{1} + n_{2}}{1 + {\left( {z^{- 1} - 1} \right)\left( {h_{11} + h_{12}} \right)}} +} \in {\cdot \frac{n_{1} - n_{2}}{1 + {\left( {z^{- 1} - 1} \right)\left( {h_{11} - h_{12}} \right)}}}} & (1)\end{matrix}$

in which n₁ and n₂ are the quantization noise from quantizers 402 a and402 b and ∈ is the mismatch between the outputs of DACs 403 a and 403 binto output summer 404. Therefore, the sum of the transfer functions(h₁₁+h₁₂) sets the global noise shaping and the difference of thetransfer functions (h₁₁−h₁₂) sets the local noise shaping of themismatch ∈.

Global noise shaping is the ability of delta-sigma modulator topology400 to shape the total noise spectrum (i.e. n₁+n₂) output fromquantizers 402 a and 402 b and summed by summer 404. In other words, theglobal noise shaping function characterizes the noise transfer function(NTF) of the total output noise, including noise attenuation in thesignal band and out-of-band noise gain. Local noise shaping is theability of delta-sigma modulator topology 400 to shape the difference inthe noise spectrums output from quantizers 402 a and 402 b into summer404 (i.e. n₁−n₂). By shaping the difference in noise spectrums n₁ andn₂, the noise demodulated by any mismatch from the outputs of DACs 403 aand 403 b into summer 404 is also shaped.

By selecting the transfer functions h₁₁ and h₁₂, and consequentlytransfer functions h₂₂ and h₂₁, a design trade-off is made between theglobal noise shaping defined by the first term of Equation 1 and thelocal noise shaping defined by the second term of Equation 1.Specifically, the total quantization noise (n₁+n₂) gain is proportionalto h₁₁+h₁₂ and the gain of the quantization noise difference (n₁−n₂)exposed to the mismatch ∈ is proportional to h₁₁−h₁₂. Therefore, todecrease the exposure to the mismatch, the term h₁₁−h₁₂, which shapesthe difference n₁−n₂ between noise spectrums, is made smaller, forexample by reducing the contribution of transfer functions h₁₂ and h₂₁.However, if the contribution of transfer functions h₁₂ and h₂₁ arereduced, the global noise shaping in the passband is also reduced inaccordance with the first term of Equation 1. Similarly, an improvementin global noise shaping by increasing the sum of the transfer functions(i.e., h₁₁+h₁₂) will generally reduce the local noise shaping capabilityby the difference of the transfer functions (i.e., h₁₁+h₁₂).

FIG. 5 is a block diagram of a representative mixedfeed-forward/feed-back delta-sigma modulator topology 500 utilizingglobal and local sets of filter stages and multiple quantizers.Modulator topology 500 may be implemented in the analog domain, for usein modulator 103 of ADC system 100 or in the digital domain for use inmodulator 114 of DAC system 111.

In modulator topology 500, the global noise shaping circuitry (filter)is shown generally by the shared modulator circuitry 501. In thisexample, two integrator stages 502 a and 502 b are shown forillustration. However, as with each of the modulator topologiesdescribed herein, the number and type of shared filter stages may varydepending on the desired noise global shaping (e.g. the number andfrequencies of the poles and zeros of the noise transfer function). Theoutputs from filter stages 502 a and 502 b are fed-forward withweighting coefficients C1 and C2 through feed-forward paths 503 a and503 b into output summer 504. Feed-forward paths 503 a and 503 b includeanalog attenuators or gain stages for analog embodiments or multipliersin digital embodiments which apply the weighting coefficient C1 and C2.A feedback path 505, with gain g1 and delay z⁻¹ and input summer 506move the zeros of the noise transfer function (“NTF”) defined by filterstages 502 a and 502 b along the unit circle in the z-plane.

The output from shared filter 501 is passed to each of n+1 number ofparallel independent feed-forward noise shaping circuits (filters) driveoutputs D0-Dn. Two exemplary independent noise shaping circuits 507 aand 507 b are shown generally in FIG. 5 for illustrative purposes. Inthis example, independent feed-forward noise shapers 507 a and 507 b arebased on two filter (integrator) stages 508 a and 508 b andcorresponding feed-forward paths 509 a and 509 b weighting theintegrator outputs by coefficients C3 and C4 into an output summer 510.As with each of the modulator topologies described herein, the numberand type of filter stages used in the independent filter sections of thegiven loop filter will vary depending on the desired local noise shapingresponse. A feedback path 511 with gain g2 and delay z⁻¹ and inputsummer 512 set the zeros of local noise transfer function.

Each independent noise shaper 507 a and 507 b includes an independentquantizer 513, which may be either a single-bit or a multiple-bitdesign. If a single-bit design is used, optional DEM circuits 107 a and107 b or 120 a and 120 b (see respective FIGS. 1A and 1B) may beeliminated from the converter. If a multiple-bit design is used, thensome minimal optional DEM circuits 107/120 are included in respectivesystems 100 and 111 (see respective FIGS. 1A and 1B) to noise shape anyremaining mismatch noise.

The quantized output from each local quantizer 513 is fed-back to thecorresponding local input summer 512 to close the independent feedbackloop. The quantized outputs of all independent quantizers 513 are summedin summer 514 and then fed-back to the inverting input of input summer515 to close the overall modulator feedback loop.

A dither source 516 is provided at the input of each local quantizer 516to reduce or eliminate tonality in the modulator outputs D0-Dn.Preferably, the amount of dither input into each quantizer 513 isselected such that the sum of the dither at the modulator outputs fromall quantizers 513 is a constant. Consequently, tonality is avoided inthe output without substantially increasing the overall noise floor.This result is accomplished by using a pseudo-random number generator(PRN) which generates bits that increase the quantized level at someoutputs and decrease the quantized level at other outputs. An exemplarydither generation source is discussed below in conjunction with FIG. 10.

The overall modulator loop including the shared global noise shapingcircuitry 501, modulator input summer 515 and the parallel independentnoise shapers 507 sets the overall noise shaping of the modulatoroutput. Each independent noise shaper 507 individually shapes the noiseto its corresponding output. However, a tradeoff generally must be madesince more feedback in each independent noise shaper 507 improves thelocal mismatch shaping but worsen the overall noise shapingcharacteristics of the modulator. Loop timing is ensured by delays 517.

FIG. 6 illustrates an exemplary feedback modulator topology 600 withboth shared and independent filters and multiple quantizers according tothe inventive principles. In this case, shared filter 601 provides themodulator front-end, and a set of n+1 number of independent filters 602a and 602 b drive the outputs D0-Dn, in which n is an integer of one orgreater. Two representative independent filters 602 a and 602 b areshown in FIG. 6 for discussion purposes. The sum of all the quantizedoutputs from independent filters 602 is generated by summer 603 and isfed-back to shared filter 602 to control the overall noise shapingcharacteristics of the modulator. As will be discussed further below,each independent filters 602 a and 602 b has its own independentfeedback loop for setting the local noise shaping characteristics of theassociated output D0-Dn.

For illustrative purposes, shared filter 601 is shown as a two-stagefeedback noise shaper including a pair of filter stages 604 a and 604 b.The number of shared filter stages 604 a and 604 b, as well as theirtransfer functions, may vary from application to application dependingon the desired NTF. In this example, two integrator stages 604 a and 604b are shown.

The feedback summed by summer 603 is weighted into summers 605 a and 605b of shared filter 601 with feedback coefficients C1 and C2.Coefficients C1 and C2 set two poles in the overall NTF at eachmodulator output. The associated NTF zeros are set on the unit circle inthe z-plane by local feedback loop (resonator) 606 having a gain g2 anda delay z⁻¹.

Each independent filter 602 a and 602 b, in conjunction with sharedfilter 601 operates as a separate noise shaper. In FIG. 6, eachindependent filter (noise shaper) 602 a-602 b is based on a pair offilter stages 607 a and 607 b. For illustration purposes, filter stages607 a-607 b are shown as integrator stages, although the number offilter stages 607 as well as their transfer functions, may vary inalternate embodiments.

Each independent filters 602 a-602 b includes a dedicated quantizer 608and a local feedback loop implemented with summers 609 a and 609 b.Coefficients C3 and C4 set two more poles of the NTF for thecorresponding output D0-Dn. A local feedback loop (resonator) 610, withgain g2 and delay z⁻¹, sets two more zeros of the NTF. Quantizer 608 iseither of a single-bit or a multiple-bit design. If the multiple-bitdesign is selected, optional DEM 107 a and 107 b or 120 a and 120 b (seerespective FIGS. 1A and 1B) is preferably utilized in system 100/111 toaddress any mismatch. Dither sources 611, as discussed further belowwith respects to FIG. 10, at the input to each quantizer 608 reduce oreliminate the tonality of the quantizer outputs. Similar to dithersources 516, if the sum of the dither added to modulator 600 is aconstant, the overall noise floor will not substantially increase. Eachfeedback loop also includes a delay 612 at the quantizer 608 output.

The shared filter section according to the present principles isalternatively disposed between the independent filters and the multiplequantizers, as demonstrated by exemplary delta-sigma modulator topology700 shown in FIG. 7. Delta-sigma modulator topology 700 includes ashared filter section 701, which has an the input driven by the sum ofthe feed-forward contributions of n+1 number of independent filters 702generated by summer 703, in which n is an integer of one or greater. Tworepresentative independent filters 702 a and 702 b are shown in FIG. 7.The output from shared filter 701 drives n+1 number of output stages,two of which are shown at 704 a and 704 b for illustrative purposes.Output stages 704 a and 704 b receive the weighed feedforward output (i₀and i_(n)) from independent filters 702 a and 702 b, respectively. Inturn, the quantized output from output stages 704 a and 704 b (FB₀ andFB_(n)) is fed-back to the input of independent filters 702 a and 702 b,respectively.

In this example, shared filter section 701 is based on a pair ofintegrator stages 705 a and 705 b having outputs fed-forward into asummer 706 through weighting stages 707 a-707 b (which includeamplifiers, multipliers or attenuators) with weighting coefficients C1and C2. Filter section 701 sets two pole-zero pairs in the global NTF.Filter 701 section is also shown with a feedback loop 708, with gain g1and delay and input summer 709 for moving the global zeros along theunit circle of the z-plane.

Independent filters 702 a-702 b are represented by an integrator stage710 and an input summer 711 summing the input signal with feedbackFB₀-FB_(n) from the output stages 704 a and 704 b. The number of stagesin independent filters 702 a and 702 b and their transfer functions mayvary depending on the application.

Output stages 704 a-704 b each include a summer 712, weighting stage 713having a weighting coefficient C2 for weighting the signal fed-forwardfrom the corresponding input filters 702 a and 702 b. A single- ormultiple-bit quantizer 714 through a delay 716 generates the outputD0-Dn. A dither source 715, similar to those already described and shownin FIG. 10, reduces or eliminate tonality at the corresponding outputD0-Dn.

A further exemplary delta-sigma modulator topology 800 embodying theprinciples of the present invention is shown in FIG. 8. Modulatortopology 800 includes a shared filter 801, n+1 number of independentfilters represented by exemplary independent filters 802 a-802 b, and asummer 803 summing the outputs from the independent filters 802 a and802 b. The feedback sum generated by modulator summer 803 is fed-back toboth shared filter 801 and each representative independent filter 802 aand 802 b. Independent filters 802 a-802 b also have local feedbackloops that will be discussed further below.

In modulator topology 800, shared filter 801 includes a pair ofintegrator stages 804 a and 804 b and feedback summers 805 a-805 c forweighed feedback of the sum from modulator summer 803 with feed-backcoefficients C1-C3. The pair of integer stages 804 a and 804 b andfeedback coefficients set two pole-zero pairs in the global NTF.

Independent filters 802 a-802 b in modulator topology 800 include a pairof integrator stages 806 a-806 b and a dedicated quantizer 807(multiple-bit or single-bit). A delay 810 is provided at the quantizeroutput to ensure proper signal timing. Each independent filter 802 a-802b also includes a pair of feedback paths 808 a and 808 b and summers 809a and 809 b. Independent feedback paths 808 a and 808 b apply localfeedback coefficients C4 and C5 which control the local noise shapingresponse. Summers 809 b also receive the sum of the feedback from summer803, weighted by feedback coefficient C3. Delays 810 ensure the propertiming and dither sources 811 (see FIG. 10) reduces or eliminatetonality in the output.

Noise shaper topology 900 shown in FIG. 9 integrates the DEM functioninto the modulator itself. In the illustrated embodiment, noise shaper900 utilizes four shared integrator stages 901 a-901 d and correspondingsummers 902 a-902 d in a feedback topology with feedback coefficientsC1-C4. A pair of feedback loops (resonators) 903 a and 903 b havingrespective gains of g1 and g2 and delays z⁻¹ set the shared zeros.

The independent filters in exemplary noise shaper 900 are represented bysingle integrator stages 904 a-904 c. The number of shared filter stages901 and independent filter stages 904, as well as their transferfunctions, vary from embodiment to embodiment depending on the desirednoise shaping characteristics.

The outputs from independent filter stages 904 a-904 c are fed-forwardwith weighting coefficients C21 and C22 into summers 905 a-905 d.Summers 905 a-905 d, which also receive the output from final sharedfilter stage 901 b, drive corresponding multiple quantizers 906 a-906 d.The quantized outputs OUT 0-OUT 3 from respective quantizers 906 a-906 dare passed through corresponding delays 911 a-911 d and then summed bysummer 907 to generate the feedback to shared filter stages 901 a-901 d.Additionally, the quantized outputs are also fed-back to summers 908a-908 c at the inputs to respective independent filter stages 904 a-904b.

In this example, independent feedback summer 908 a takes the differencebetween the sum of OUT 0 and OUT 1 and the sum of OUT 2 and OUT 3.Independent feedback summer 908 b takes the difference between OUT 0 andOUT 1 and feedback summer 908 c takes the difference between OUT 2 andOUT 3. After filtering by independent filter stages 904 a-904 c, summers905 a and 905 b take the following weighted combinations. Summer 905 atakes the difference between the output of shared filter 901 d, theoutput of independent filter 904 a, and the output of independent filter904 b. Summer 905 b subtracts the output of independent filter 904 afrom, and adds the output of independent filter 904 b to, the output ofshared filter 901 d. Summer 905 c subtracts the output of independentfilter 904 c from, and adds the output of independent filter 904 a to,the output of shared filter 901 d. Finally, summer 905 d takes the sumof the output of shared filter 901 d; the output of independent filter904 a and the output of independent filter 904 c. Four dither sources910 a-910 d are summed into the 4 quantizers to help eliminate patterns;the total dither applied always sums to a constant. In a typicaldelta-sigma modulator, this dithering would increase the noise floor. Inthis topology, it is possible to increase the noise less by having thedither sources sum to a constant.

FIG. 10 is a schematic representation of an exemplary dither source(generator) 1000 suitable for generating dither streams which add to aconstant. Dither generator includes a pseudo-random number generator(PRN) 1001 and a set of summers 1002 a-1002 d for a four-bit system. Inthis example, the pseudo-random number streams n1-n4 and the complementsare summed in the following manner to generate dither streams d1-d4:

d 1=n 1+n 2

d 2=n 4−n 2

d 3=−n 1+n 3

d 4=−n 4−n 3

In this example, dither d1+d2+d3+d4 sum to a constant of zero (0). Thisimplementation would be especially appropriate to D/A converters, as thefilters 904 a, b, c, are very simple having only integer inputs and maybe implemented with very few bits of memory.

In each of the topology described above, appropriate selection of theindividual shared and independent filter stages result in a veryefficient design. Generally, since the earlier stages in the loop filterrequire more accuracy, the earlier stages are made larger and withincreased fabrication accuracy. In turn, the later filter stages in theloop are made smaller and with less fabrication accuracy. Moreover, ifthe unique stages in the loop filter are placed later in the loop, thoseunique stages are less accurate without adversely impacting the overallmodulator performance.

Although the invention has been described with reference to specificembodiments, these descriptions are not meant to be construed in alimiting sense. Various modifications of the disclosed embodiments, aswell as alternative embodiments of the invention will become apparent topersons skilled in the art upon reference to the description of theinvention. It should be appreciated by those skilled in the art that theconception and the specific embodiment disclosed may be readily utilizedas a basis for modifying or designing other structures for carrying outthe same purposes of the present invention. It should also be realizedby those skilled in the art that such equivalent constructions do notdepart from the spirit and scope of the invention as set forth in theappended claims.

It is therefore, contemplated that the claims will cover any suchmodifications or embodiments that fall within the true scope of theinvention.

What is claimed is:
 1. A noise shaper comprising: first and secondquantizers; first and second feedback paths each providing feedback froma corresponding quantizer output; and a loop filter system implementinga plurality of functions including a first non-zero transfer functionbetween the first feedback path and an input the first quantizer, asecond non-zero transfer function between the first feedback path and aninput of the second quantizer, a third non-zero transfer functionbetween the second feedback path and the input of the first quantizerand a fourth non-zero transfer function between the second feedback pathand the input the second quantizer.
 2. The noise shaper of claim 1wherein the first transfer function is approximately equivalent to thefourth transfer function and the second transfer function isapproximately equivalent to the third transfer function.
 3. The noiseshaper of claim 1 wherein the plurality of quantizers comprises n numberof quantizers, the plurality of feedback loops comprises n number offeedback loops and the loop filter system implements n² number oftransfer functions between outputs of each of the n number of feedbackloops and inputs of the n number of quantizers.
 4. A noise shapercomprising: a loop filter including a shared filter and first and secondindependent filters; a first quantizer for generating a first quantizedoutput signal from a first output of the loop filter, the firstquantized output signal fed-back to the first independent filter; and asecond quantizer for generating a second quantized output signal from asecond output of the loop filter, the second quantized output signalfed-back to the second independent filter.
 5. The noise shaper of claim4 further comprising a feedback loop for feeding-back a sum of the firstand second quantized output signals to the shared filter.
 6. The noiseshaper of claim 4 wherein an output signal from the shared filter drivesinputs of the first and second independent filters.
 7. The noise shaperof claim 4 wherein output signals from the first and second independentfilters drive an input of the shared filter.
 8. The noise shaper ofclaim 4 further comprising dither generation circuitry for generatingdither at inputs of the first and second quantizers.
 9. The noise shaperof claim 8 wherein the dither at the inputs of the first and secondquantizers are non-equal and sum to a constant at corresponding firstand second outputs of the modulator.
 10. The noise shaper of claim 4wherein at least one of the shared and independent filters has afeedforward topology.
 11. The noise shaper of claim 4 wherein at leastone of the shared and independent filters has a feedback topology. 12.The noise shaper of claim 4 wherein a selected one of the quantizerscomprises a single-bit quantizer.
 13. The noise shaper of claim 4wherein a selected one of the quantizers comprises a multiple-bitquantizer.
 14. A delta-sigma modulator comprising: a plurality ofquantizers for generating a plurality of quantized output signals inresponse to outputs from a loop filter; and a loop filter driving inputsof the plurality of quantizers, the loop filter including at least oneshared filter stage receiving a sum of feedback from each of theplurality of quantizers for characterizing a global modulator noisetransfer function of the quantized output signals and a plurality ofindependent filter stages each receiving feedback from a correspondingone of the quantizers for characterizing a local noise transfer functionof a corresponding one of the quantized output signals.
 15. Thedelta-sigma modulator of claim 14 wherein the shared filter receives amodulator input signal and drives inputs to the plurality of independentfilter stages.
 16. The delta-sigma modulator of claim 14 wherein atleast one of the plurality of independent filters receives a modulatorinput signal and drives an input to the shared filter.
 17. Thedelta-sigma modulator of claim 14 further comprising a dither source forproviding dither at an input of each of the plurality of quantizers. 18.The delta-sigma modulator of claim 14 wherein the dither provided to theinputs of the plurality of quantizers sums to a constant value.
 19. Thedelta-sigma modulator of claim 14 wherein the plurality of independentfilters receive a modulator input signal and drive an input of theshared filter and an output of the shared filter drives inputs of theplurality of quantizers.
 20. The delta-sigma modulator of claim 19wherein the plurality of independent filters each feed-forward afeedforward signal to the input of a corresponding one of thequantizers.
 21. The delta-sigma modulator of claim 14 wherein at leastone of independent filters receives a feedback signal from a sum of thequantized output signals.
 22. A method of noise shaping comprising:characterizing local noise shaping of each of first and second signalswith corresponding first and second independent filter sets each havingat least one filter stage; characterizing global noise shaping of thefirst and second signals with a shared filter set having at least onefilter stage shared by the first and second sets of independent filterstages; and independently quantizing the first and second signals withfirst and second quantizers.
 23. The method of claim 22 furthercomprising: feeding-back a first quantized signal from the firstquantizer to the first independent filter set; feeding-back a secondquantized signal from the second quantizer to the second independentfilter set; and feeding-back a sum of the first and second quantizedsignals to the shared filter set.
 24. The method of claim 22 furthercomprising applying dither at inputs of the first and second quantizersto reduce tonality in the first and second quantized output signals. 25.The method of claim 24 wherein said step of applying dither at theinputs of the first and second quantizers further comprises selectivelyapplying dither at the inputs of the first and second quantizers whichsum to a constant.
 26. The method of claim 22 further comprising drivinginputs of the independent filter sets with an output of the sharedfilter set.
 27. The method of claim 22 further comprising driving aninput of the shared filter set with an output of at least one of theindependent filter sets.
 28. A data conversion system for convertingdata from a first form to a second form comprising: a delta-sigmamodulator for modulating an input signal comprising: a loop filterincluding a shared filter stage and a plurality of independent filterstages each sharing a signal path with the shared filter stage; and aplurality of quantizers for generating independently quantized outputsignals from outputs of the loop filter, an output of each quantizerfed-back to a corresponding one of the independent filter stages; and adigital to analog converter for converting the quantized outputs fromthe plurality of quantizers into analog.
 29. The data conversion systemof claim 28 wherein a sum of the quantized outputs from the plurality ofquantizers is fed-back to the shared filter stage.
 30. The dataconversion system of claim 28 wherein the system converts data fromdigital to analog form and the digital to analog converter comprises aoutput digital to analog converter.
 31. The data conversion system ofclaim 28 wherein the system converts data from analog to digital formand the digital to analog converter comprises a feedback digital toanalog converter.
 32. The data conversion system of claim 28 and whereinthe plurality of quantizers comprise single-bit quantizers.
 33. The dataconversion system of claim 28 wherein the plurality of quantizerscomprise multiple-bit quantizers and the system further comprisesdynamic element matching circuitry between outputs of the plurality ofmultiple-bit quantizers and the digital to analog converter.
 34. A dataconversion system comprising a delta-sigma modulator with multiplequantizers and multiple dither noise sources, wherein the sum of thedither noise sources has significantly less power than the sum of thepowers of the noise sources.
 35. The data conversion system of claim 34wherein the sum of the dither noise sources is approximately zero.
 36. Amethod of noise shaping comprising: filtering an input signal with ashared filter contributing to the inputs of each of a plurality ofquantizers, a sum of outputs from the quantizers fed-back to an input ofthe shared filter to globally noise shape a sum of the spectrums of theoutputs of the quantizers; and filtering a difference between theoutputs of the quantizers with a local filter system, the filtereddifference fed-back to inputs of each quantizer to locally noise shape adifference of the spectrums of the outputs of the quantizers.
 37. Themethod of claim 36 wherein filtering the difference comprises: taking adifference between outputs of first and second quantizers; and filteringthe difference with a local filter system comprising a shared localfilter.
 38. The method of claim 36 wherein filtering the differencecomprises: filtering an output of the shared filter with a local filtersystem comprising a plurality of parallel filters, each parallel filterreceiving feedback from a corresponding one of the quantizers.
 39. Themethod of claim 36 wherein filtering the difference comprises filteringthe spectrum of the difference of the output of the quantizers to shapemismatch noise in a set of conversion elements.